Phase-adjusted channel estimation for frequency division multiplexed channels

ABSTRACT

A method and apparatus for estimating a frequency response of a channel. The method includes adjusting phase components of estimates of the frequency response to provide phase-adjusted estimates; performing a smoothing operation on the phase-adjusted estimates to provide smoothed phase-adjusted estimates; and generating an output of a reverse phase adjustment, wherein the reverse phase adjustment is performed on the smoothed phase-adjusted estimates.

CROSS-REFERENCE TO RELATED APPLICATIONS

This present disclosure is a continuation of U.S. application Ser. No.11/483,057, filed on Jul. 6, 2006, which claims priority under 35 U.S.C.§119(e) to U.S. Provisional Application No. 60/727,815, filed Oct. 18,2005, and 60/799,922, filed May 12, 2006.

BACKGROUND OF THE INVENTION

This invention relates to channel estimation, and, more particularly, tousing phase adjustment in estimating the frequency response of afrequency division multiplexed communication channel.

The concept of a digital communications channel is well known. Inparticular, it is known that a channel can affect the amplitude andphase of a signal carried by the channel. As a simple example, suppose asignal cos(ω₀t′) is communicated in a channel, where w₀ is the angularfrequency of the signal, t′ is the time associated with the transmissionof the signal, and t′=0 represents the beginning of the transmission.Ideally, the signal that arrives at a receiver should have the sameamplitude and phase; i.e., the received signal should be cos(ω₀t), wheret is the time associated with receiving the signal, and t=0 representsthe beginning of the reception. The time t=0 may correspond to t′=τ, forsome transmission delay τ. However, the received signal is seldom thesame as the transmitted signal, even in a noiseless environment. Rather,(in the absence of noise) a receiver will more likely receive a signal Acos(ω₀t+φ), where A is a real number that shows the channel's effect onthe amplitude of the signal, and φ is a real number that shows thechannel's effect on the phase of the signal. The quantity φ is commonlyreferred to as “initial phase.”

Although the example above shows a transmission signal that has only onefrequency component ω=ω₀, a signal may include more than one frequencycomponent. Additionally, a channel may affect each frequency componentdifferently. Accordingly, the amplitude A and the initial phase φ in theexample above may only apply to frequency component ω=ω₀. From thispoint on, when a signal includes more than one frequency component, theamplitude and initial phase for each frequency component ω=ω_(i) will bedenoted with a corresponding subscript i.

A fundamental concept of digital communications is that amplitude andinitial phase can be represented by a coordinate in a Cartesian plane.For example, an amplitude A and an initial phase φ can be represented bythe coordinate (x,y) where x=A cos(φ) and y=A sin(φ). Conversely, givena coordinate (x,y), an amplitude and initial phase can be computed byA=√{square root over (x²+y²)} and φ=arctan(y/x). Another fundamentalconcept is that a coordinate (x,y) can also correspond to a complexnumber of the form (x+jy), where j is the imaginary unit. In this case,the x-axis represents the real part of a complex number, and the y-axisrepresents the imaginary part of a complex number. The benefits ofrepresenting amplitude and initial phase graphically as a coordinatepoint and mathematically as a complex number are that theserepresentations allow changes in amplitude and initial phase to beeasily illustrated and computed. The next paragraph shows an example ofcomputing a channel's effects on a signal's amplitude and initial phase.In particular, an important computation involves Euler's formula, whichstates that a complex number (x+jy) can equivalently be expressed asAe^(jφ), where, as shown above, A=√{square root over (x²−y²)} andφ=arctan(y/x).

As an example, suppose a transmitted signal in a channel has frequencycomponents of the form A_(i) cos(ω_(i)t+φ_(i)). In the absence of noise,the channel will generally alter the amplitude multiplicatively by afactor K_(i), and alter the initial phase additively by a factor θ_(i),resulting in a received frequency component of the form K_(i)A_(i)cos(ω_(i)t+φ_(i)+θ_(i)). Representing these amplitudes and initialphases mathematically, the amplitude and initial phase of thetransmitted frequency components can be characterized by A_(i)e^(jφ)^(i) , and those of the received frequency component can becharacterized by K_(i)A_(i)e^(j(φ) ^(i) ^(+θ) ^(i) ⁾=A_(i)e^(jφ) ^(i)K_(i)e^(jθ) ^(i) . This shows two important things. First, it can beseen that the channel's effect on the amplitude and initial phase of thetransmitted frequency component is captured by the term K_(i)e^(jθ) ^(i). Second, if (in the absence of noise) the amplitude and initial phaseof a received frequency component is B_(i)e^(jφ) ^(i) , then thechannel's effect on the transmitted amplitude and initial phase can becomputed by

${{{K_{i}{\mathbb{e}}^{j\;\theta_{1}}} = {\frac{B_{i}{\mathbb{e}}^{j\;\varphi_{1}}}{A_{i}{\mathbb{e}}^{j\;\phi_{1}}} = {\frac{B_{i}}{A_{i}}{\mathbb{e}}^{j{({\varphi_{1} - \phi_{i}})}}}}};{i.e.}},{K_{i} = {{\frac{B_{i}}{A_{i}}\mspace{14mu}{and}\mspace{14mu}\theta_{i}} = {\varphi_{i} - {\phi_{i}.}}}}$When all of the effects K_(i)e^(jθ) ^(i) across a continuous frequencyrange are quantified, the result is a function showing a channel'seffect on signal amplitude and initial phase based on frequency. Thefunction is referred to in the art as a “transfer function.” A graph ofa transfer function with respect to frequency is referred to as thechannel's “frequency response.”

The examples above assume an absence of noise in or affecting thechannel. As mentioned above, a signal's frequency component can haveamplitude and initial phase that are represented by a complex numbers_(i), and the channel's frequency response for the frequency componentcan be represented by a complex number h_(i). In the absence of noise,the received frequency component has amplitude and initial phase givenby y_(i)=h_(i)·s_(i). However, a channel's frequency response can varyover time. Therefore, the value of h_(i) may need to be re-evaluated.One way in which this can be accomplished is by sending the receiver a“training signal,” which is a predetermined signal that is known by thereceiver. The training signal can include a frequency component that haspredetermined amplitude and initial phase given by s_(i). When thetraining signal arrives at the receiver with amplitude and initial phasey_(i), the receiver can evaluate the channel's frequency response forthe frequency component by computing h_(i)=y_(i)/s_(i).

However, when noise is present, the channel's frequency response becomesmore difficult to estimate. As used herein, the term “noise” refers tophenomena or effects, in or affecting a channel, that affect a signalcarried on the channel and that are not already included by thechannel's frequency response. Generally, when noise is present, thereceived frequency component becomes y_(i)=h_(i)·s_(i)+n_(i). In thissituation, both h_(i) and n_(i) may vary over time, and it becomes moredifficult to estimate the frequency response h_(i) with certainty basedon knowing only the training component s_(i) and the received componenty_(i).

Ultimately, the desired operation of a receiver is to correctly detect atransmitted signal. To do so, a receiver can benefit from having a moreaccurate estimate of the channel's frequency response. Additionally, achannel estimate is useful for many kinds of operations, such asequalization. However, the presence of noise undermines the receiver'sability to produce an accurate channel estimate. Accordingly, there iscontinued interest in improving a receiver's channel estimationcapabilities.

SUMMARY OF THE INVENTION

The disclosed technology provides an apparatus and method for estimatingthe frequency response of a frequency division multiplexed (FDM) channelthat includes sub-channels.

In accordance with one aspect of the invention, the disclosed technologycan compute initial estimates of the frequency response of thesub-channels based on a received signal and a predetermined trainingsignal, wherein the received signal corresponds to the predeterminedsignal and wherein the initial estimates include phase components,adjust the phase components of the initial estimates to providephase-adjusted estimates, perform a smoothing operation on thephase-adjusted estimates to provide smoothed estimates, and perform areverse phase adjustment on the smoothed estimates to provide finalestimates of the frequency response of the sub-channels.

In one aspect of the invention, for each of the sub-channels, thedisclosed technology can compute the initial estimate for thesub-channel based on portions of the received signal and thepredetermined signal that correspond to the sub-channel.

In one aspect of the invention, for each of at least one of thesub-channels, the disclosed technology can adjust the phase componentsof the initial estimates by computing a phase difference between thephase component of the initial estimate for the sub-channel and thephase component of the initial estimate for an adjacent sub-channel. Thedisclosed technology can compute a mean phase difference based on thephase differences and can adjust the phase components of the initialestimates based on the mean phase difference to provide thephase-adjusted estimates.

In one embodiment, the sub-channels include at least a first sub-channelthat corresponds to a first frequency and a second sub-channel thatcorresponds to a second frequency. The disclosed technology can adjustthe phase components of the initial estimates based on the mean phasedifference by adjusting the phase component of the initial estimate forthe first sub-channel by a first multiple of the mean phase difference,and adjusting the phase component of the initial estimate for the secondsub-channel by a second multiple of the mean phase difference. In oneembodiment, the first multiple is greater than the second multiple ifthe first frequency is greater than the second frequency, and the firstmultiple is less than the second multiple if the first frequency is lessthan the second frequency. In one embodiment, the first multiple differsfrom the second multiple by an amount that is proportional to adifference between the first and second frequency values.

In one aspect of the invention, for each sub-channel, the disclosedtechnology can adjust a phase component of the smoothed estimate by anamount that is substantially the opposite of the amount of phaseadjustment to the initial estimate for the sub-channel.

In one aspect of the invention, for each of the sub-channels, thedisclosed technology can compute the smoothed estimate for thesub-channel based on the phase-adjusted estimate for the sub-channel andat least the phase-adjusted estimate for an adjacent sub-channel. In oneembodiment, the disclosed technology can compute an average of thephase-adjusted estimate for the sub-channel and the phase-adjustedestimates for a particular number of neighboring sub-channel.

In one aspect of the invention, the disclosed technology can access anindicator that has an on-value and at least one other value. The phaseadjustment operation, the smoothing operation, and the reverse phaseadjustment are performed only when the indicator has the on-value.

In accordance with one aspect of the invention, a receiver can includemeans for computing initial estimates of the frequency response ofsub-channels in a FDM channel based on a received signal and apredetermined training signal, wherein the received signal correspondsto the predetermined signal and wherein the initial estimates includephase components, means for adjusting the phase components of theinitial estimates to provide phase-adjusted estimates, means forperforming a smoothing operation on the phase-adjusted estimates toprovide smoothed estimates, and means for performing a reverse phaseadjustment on the smoothed estimates to provide final estimates of thefrequency response of the sub-channels.

In one aspect of the invention, for each of the sub-channels, thereceiver can include means for computing the initial estimate for thesub-channel based on portions of the received signal and thepredetermined signal that correspond to the sub-channel.

In one aspect of the invention, for each of at least one of thesub-channels, the receiver can include means for adjusting the phasecomponents of the initial estimates by computing a phase differencebetween the phase component of the initial estimate for the sub-channeland the phase component of the initial estimate for an adjacentsub-channel, means for computing a mean phase difference based on thephase differences, and means for adjusting the phase components of theinitial estimates based on the mean phase difference to provide thephase-adjusted estimates.

In one embodiment, the sub-channels include at least a first sub-channelthat corresponds to a first frequency and a second sub-channel thatcorresponds to a second frequency. The receiver can include means foradjusting the phase components of the initial estimates based on themean phase difference, which can include means for adjusting the phasecomponent of the initial estimate for the first sub-channel by a firstmultiple of the mean phase difference, and means for adjusting the phasecomponent of the initial estimate for the second sub-channel by a secondmultiple of the mean phase difference. In one embodiment, the firstmultiple is greater than the second multiple if the first frequency isgreater than the second frequency, and the first multiple is less thanthe second multiple if the first frequency is less than the secondfrequency. In one embodiment, the first multiple differs from the secondmultiple by an amount that is proportional to a difference between thefirst and second frequency values.

In one aspect of the invention, for each sub-channel, the receiver caninclude means for adjusting a phase component of the smoothed estimateby an amount that is substantially the opposite of the amount of phaseadjustment to the initial estimate for the sub-channel.

In one aspect of the invention, for each of the sub-channels, thereceiver can include means for computing the smoothed estimate for thesub-channel based on the phase-adjusted estimate for the sub-channel andat least the phase-adjusted estimate for an adjacent sub-channel. In oneembodiment, the receiver can include means for computing an average ofthe phase-adjusted estimate for the sub-channel and the phase-adjustedestimates for a particular number of neighboring sub-channel.

In one aspect of the invention, the receiver can include means foraccessing an indicator that has an on-value and at least one othervalue. In one embodiment, the phase adjustment means, the smoothingoperation means, and the reverse phase adjustment means can performtheir operations only when the indicator has the on-value.

In accordance with one aspect of the invention, the receiver can includea computer program running on a processor that computes initialestimates of the frequency response of sub-channels in a FDM channelbased on a received signal and a predetermined training signal, whereinthe received signal corresponds to the predetermined signal and whereinthe initial estimates include phase components, adjusts the phasecomponents of the initial estimates to provide phase-adjusted estimates,performs a smoothing operation on the phase-adjusted estimates toprovide smoothed estimates, and performs a reverse phase adjustment onthe smoothed estimates to provide final estimates of the frequencyresponse of the sub-channels.

In one aspect of the invention, for each of the sub-channels, a computerprogram running on a processor can compute the initial estimate for thesub-channel based on portions of the received signal and thepredetermined signal that correspond to the sub-channel.

In one aspect of the invention, for each of at least one of thesub-channels, a computer program running on a processor can adjust thephase components of the initial estimates by computing a phasedifference between the phase component of the initial estimate for thesub-channel and the phase component of the initial estimate for anadjacent sub-channel. A computer program running on a processor cancompute a mean phase difference based on the phase differences and canadjust the phase components of the initial estimates based on the meanphase difference to provide the phase-adjusted estimates.

In one embodiment, the sub-channels include at least a first sub-channelthat corresponds to a first frequency and a second sub-channel thatcorresponds to a second frequency. A computer program running on aprocessor can adjust the phase components of the initial estimates basedon the mean phase difference by adjusting the phase component of theinitial estimate for the first sub-channel by a first multiple of themean phase difference, and adjusting the phase component of the initialestimate for the second sub-channel by a second multiple of the meanphase difference. In one embodiment, the first multiple is greater thanthe second multiple if the first frequency is greater than the secondfrequency, and the first multiple is less than the second multiple ifthe first frequency is less than the second frequency. In oneembodiment, the first multiple differs from the second multiple by anamount that is proportional to a difference between the first and secondfrequency values.

In one aspect of the invention, for each sub-channel, a computer programrunning on a processor can adjust a phase component of the smoothedestimate by an amount that is substantially the opposite of the amountof phase adjustment to the initial estimate for the sub-channel.

In one aspect of the invention, for each of the sub-channels, a computerprogram running on a processor can compute the smoothed estimate for thesub-channel based on the phase-adjusted estimate for the sub-channel andat least the phase-adjusted estimate for an adjacent sub-channel. In oneembodiment, a computer program running on a processor can compute anaverage of the phase-adjusted estimate for the sub-channel and thephase-adjusted estimates for a particular number of neighboringsub-channel.

In one aspect of the invention, a computer program running on aprocessor can access an indicator that has an on-value and at least oneother value. The phase adjustment operation, the smoothing operation,and the reverse phase adjustment are performed only when the indicatorhas the on-value.

Further features of the invention, its nature and various advantages,will be more apparent from the accompanying drawings and the followingdetailed description of the various embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an exemplary communication system inaccordance with aspects of the invention;

FIG. 2 is a graph of an exemplary transfer function for a frequencydivision multiplexed (FDM) channel applicable to the disclosedinvention;

FIG. 3 is a graph of an exemplary transfer function for a FDM channelapplicable to the disclosed invention;

FIG. 4 is a graph of exemplary frequency components for adjacentsub-channels of a FDM channel applicable to the disclosed invention;

FIG. 5 is a flow diagram of an exemplary method for estimating afrequency response of an FDM channel in accordance with one aspect ofthe invention;

FIG. 6A is a block diagram of an exemplary high definition televisionthat can employ the disclosed technology;

FIG. 6B is a block diagram of an exemplary vehicle that can employ thedisclosed technology;

FIG. 6C is a block diagram of an exemplary cell phone that can employthe disclosed technology;

FIG. 6D is a block diagram of an exemplary set top box that can employthe disclosed technology; and

FIG. 6E is a block diagram of an exemplary media player that can employthe disclosed technology.

DETAILED DESCRIPTION

The disclosed technology provides a method and an apparatus forestimating the frequency response of a frequency division multiplexed(FDM) channel. As used herein, frequency division multiplexing includesorthogonal FDM. As described above herein, a frequency responsedescribes a channel's effect on the amplitude and initial phase offrequency components in a signal. It is important to have an estimate ofa channel's frequency response in digital communications because digitalinformation is often represented by a signal's amplitude and/or phase.For example, this is the case in pulse amplitude modulation andquadrature amplitude modulation. An estimate of a channel's frequencyresponse can allow a receiver to reverse the effects of the channel byusing equalization. Ideally, after equalization, the only remainingeffects distorting a frequency component is noise, and the effects ofnoise can be mitigated or eliminated by channel coding. From this pointon, an estimate of a channel's frequency response may be referred tosimply as a “channel estimate.”

One use of a channel estimate is in equalization (which isolates noise),but noise is a reason why it is difficult to produce an accurate channelestimate in the first place. In relation to a channel, the conceptualdifference between frequency response and noise is clear. As describedabove, “noise” as used herein refers to phenomena or effects that affecta signal carried on a channel and that are not already included by thechannel's frequency response. Analytically, these effects can bedescribed in the frequency domain by Y(ω)=H(ω)·S(ω)+N(ω), where H(w) isthe channel frequency response, S(w) is the transmitted signal'sfrequency content, Y(w) is the received signal's frequency content, andN(w) is the noise frequency content. Given this equation, it can be seenthat H(w) and N(w) are quite distinct mathematically. However, inpractice, which effects contribute to noise and which effects contributeto frequency response may not be easily distinguishable. Thecharacterization of various effects as contributing to noise orfrequency response is not crucial, however, because, as the followingdescription will show, noise and frequency response have differentcharacteristics and properties, which can be used to estimate thefrequency response of a channel.

Referring now to FIG. 1, there is shown a block diagram of acommunication system 100 in accordance with one aspect of the invention.The communication system 100 includes a transmitter 102 and a receiver104, which can be in communication with each other by a physical path106, a wireless path 108, or a combination thereof. The physical path106 may include one or more communication media or devices (e.g., wires,cables, optical fiber, repeater devices) and, in some embodiments, mayalso include wireless segments (not shown). The wireless “path” 108 maynot be a single path and may include multipath effects, such as fading.

In accordance with one aspect of the invention, the receiver 104includes a channel estimation circuit 110, and the transmitter includesa corresponding training circuit 112. The training circuit 112 operatesto communicate a predetermined training signal (not shown) to thechannel estimation circuit 110, which also knows the predeterminedtraining signal. The channel estimation circuit 110 receives acorresponding signal (not shown), and it is assumed that any differencebetween the received signal and the predetermined training signal is aresult of noise and the channel's frequency response. The illustratedreceiver 104 also includes a control circuit 114, which may be optionalin certain embodiments. The operation of the control circuit 114 will bedescribed later herein. Additionally, the components 116-122 of thechannel estimation circuit 110 will also be described later herein.

Referring now to FIG. 2, there is shown a graph of an exemplaryfrequency division multiplexed (FDM) channel. Those skilled in the artwill recognize that a FDM channel is described in terms of frequency (f)or angular frequency (w). These terms are used interchangeably becausethey are related by the relationship a ω=2πf. From this point on, andfor convenience, the term frequency will be used to refer to theillustrations, even though the illustrations show angular frequency. Itwill be understood that the discussion of frequency also applies toangular frequency.

A FDM channel is described by a frequency band of interest, which, asused herein, refers to a continuous range of frequencies on whichinformation can be communicated. The particular frequency band used in aparticular instance may depend upon industry specifications, governmentregulations, technological considerations, and/or economicconsiderations. Those skilled in the art will recognize that FDMoperates to separate a frequency band of interest into sub-bands. FIG. 2shows a portion of a frequency band of interest that includes foursub-bands 202-208. The sub-bands may also be referred to herein as“sub-channels” or “tones.” Sub-channels that are next to each other willbe referred to herein as being “adjacent” to each other. FIG. 2 showsthat each sub-channel 202-208 includes a particular frequency w_(i).However, those skilled in the art will recognize that each sub-channel202-208 may also include other frequencies that are not expresslylabeled. The frequencies w_(i) and the sub-channels 202-208 may or maynot be uniformly spaced. From this point on, it will be assumed that thecarrier signal for each sub-channel 202-208 has the illustratedfrequency w_(i).

FIG. 2 also shows a curve 210 that represents the magnitude of thechannel's frequency response |H(w)| with respect to frequency. Thismagnitude curve 210 shows the channel's amplitude effect at differentfrequencies. In contrast, FIG. 3 shows a curve 302 that represents thechannel's phase effect for different frequencies. The terms K_(i)e^(jθ)^(i) previously used herein correspond to the magnitude and phase curves210, 302 of FIGS. 2-3. In particular, the value of K_(i) is related tothe magnitude curve 210 at ω=ω_(i), and the value of θ_(i) is related tothe phase curve 302 at ω=ω_(i). As described above, the curves 210, 302may vary over time. Computing estimates of K_(i)e^(jθ) ^(i) acrossdifferent frequencies will provide estimates of particular points in thecurves 210, 302. In one embodiment, some portions of the curves 210, 302may be estimated by interpolation. For example, the portions of thecurves between the frequencies w_(i) can be computed by interpolation.

With continuing reference to FIGS. 2-3, it can be seen that themagnitude and phase curves 210, 302 are relatively smooth and gradual.One reason for the gradual progression is that the effects and phenomenathat contribute to a frequency response generally affect neighboringfrequencies in similar ways. For example, assuming that the frequenciesw₀ and w₁ are not too far apart, an effect that contributes to thefrequency response will affect signals having frequencies w₀ and w₁ insimilar ways. One way to measure this similarity is by using amathematical concept called “correlation,” which measures therelationship between two quantities. Essentially, for two quantities xand y, correlation measures how closely the relationship between thequantities comes to x=y over time. If the two quantities rise and fallby exactly the same amount and at the same time, the quantities are saidto be “perfectly correlated.” If the quantities rise and fall bydifferent amounts and/or with a delay, then the quantities can still besaid to be correlated, but they are not perfectly correlated. In termsof correlation, it is generally the case that the amplitude effect andphase effect of adjacent sub-channels are correlated if the sub-channelsare close enough to each other. From this point on, it will be assumedthat the sub-channels in a FDM channel are close enough to each other tobe correlated. Accordingly, referring to the illustrated sub-channels202-208 of FIGS. 2-3, by way of example, the magnitude curve 210 and thephase curve 302 in adjacent sub-channels should vary in similardirections and by similar amounts.

In contrast, it is generally the case that noise in neighboringsub-channels is not correlated. Rather, noise in different sub-channelsis generally “independent,” which essentially means that there is nodiscernable relationship between the values of and changes in the noisein the sub-channels. However, there may be circumstances in which noisein different sub-channels are correlated, and these will be examined inmore detail later herein.

FIG. 4 illustrates one example of the frequency response effects andnoise effects on a predetermined training signal with amplitude andinitial phase s_(i) for different frequency components. The illustrationuses complex planes to show the amplitudes and initial phases of signalsin adjacent frequencies ω=ω_(i) and ω=ω_(i+1). For example, the complexplane on the left side of FIG. 4 can correspond to the sub-channelcontaining ω=ω₀ in FIGS. 2-3, and the complex plane on the right side ofFIG. 4 can correspond to the adjacent sub-channel containing ω=ω₁ inFIGS. 2-3. In the left sub-channel, for a frequency component withamplitude and phase s_(i), the result of the sub-charmers effect on thesignal is h_(i)·s_(i). In the right sub-channel, for a frequencycomponent with amplitude and phase s_(i+1), the result of thesub-channel's effect on the signal is h_(i+1)·s_(i+1). As mentionedabove, we are assuming that the two adjacent sub-channels arecorrelated. Therefore, the amplitude and phase effects of the left andright sub-channels should be similar. In the illustration, it is assumedthat the sub-channels are perfectly correlated. Therefore, assuming thats_(i)=s_(i+1), then h_(i)·s_(i)=h_(i+1)·s_(i+1) and both are representedby the vector 402. The use of vectors is a convenient way to illustratethe changes in amplitude and phase.

As described above, noise in the two sub-channels are generallyindependent. As an example of this independence, FIG. 4 illustrates thenoise vector n_(i) in the left sub-channel and the noise vector n_(i+1)in the right sub-channel as having different magnitudes and directions.Therefore, in the illustration, the noise in the two sub-channelsaffects their frequency components differently. The result of the noiseeffect in the left sub-channel is the amplitude and phasey_(i)=h_(i)·s_(i)+n_(i), shown by vector 404, and the result of thenoise effect in the right sub-channel is the amplitude and phasey_(i+1)=h_(i+1)·s_(i+1)+n_(i+1), shown by vector 406. It can be seenthat the noise effect alone has caused the resulting vectors y_(i) andy_(i+1) to be quite different. Therefore, the results of estimatingh_(i) based on y_(i) alone and estimating h_(i+1) based on y_(i+1) alonewill be different, even though they should be the same in theillustrated example.

Returning now to FIG. 1, the channel estimation circuit 110 operates toestimate the frequency response of the channel from the transmitter 102to the receiver 104. The transmitter's training circuit 112 can send apredetermined training signal to the receiver 104, and the receiver 104can receive a signal corresponding to the predetermined signal. Thepredetermined signal can have predetermined amplitudes and phases s_(i)for frequency components w_(i) in the sub-channels of the FDM channel.The corresponding amplitudes and phases in the frequency components ofthe received signal are denoted as y_(i). The channel estimation circuit110 includes an initial estimation circuit 116 that can compute initialestimates of the frequency response. In one embodiment, the initialestimates can be obtained by computing the result of the operationsĥ_(i)=y_(i)/s_(i). It can be seen that, analytically,ĥ_(i)=h_(i)+n_(i)/s_(i), and the only difference between the initialestimate and the actual frequency response is due to noise.

In accordance with one aspect of the invention, the channel estimationcircuit 110 makes use of the realization that the frequency response ofadjacent sub-channels are correlated, but the noise effects in thesub-channels are generally less correlated or not correlated. Based onthe correlation between the channel effects of adjacent sub-channels,h_(i) and h_(i+1) should be similar. Therefore, averaging the frequencyresponses of the adjacent sub-channels should approximate the frequencyresponses of the sub-channels. In contrast, based on the lesser ornon-correlation of noise between sub-channels, n_(i) and n_(i+1) arelikely to be different. Therefore, averaging the noise effects of theadjacent sub-channels may mitigate the noise effects. The smoothingcircuit 120 operates to perform this averaging operation. In oneembodiment, if the averaging operation involves two sub-channels, theoperation produces

$\frac{{\hat{h}}_{i} + {\hat{h}}_{i + 1}}{2} \approx {h_{i} + {\frac{n_{i} + n_{i + 1}}{2s_{i}}.}}$If n_(i)=n_(i+1), the averaging operation does not improve the initialestimate ĥ_(i). At best, n_(i)=−n_(i+1) and the noise effects cancelout, leaving

$\frac{{\hat{h}}_{i} + {\hat{h}}_{i + 1}}{2} \approx {h_{i}.}$In certain circumstances, the averaging operation may exacerbate thechannel estimates for certain frequencies. For example, in the case that|n_(i+1)|>|n_(i)|, the averaged estimate is actually worse than theinitial estimate. Also, if arg

${\left( \frac{n_{i}}{s_{i}} \right) \approx {\pm \mspace{11mu}{\arg\left( h_{i} \right)}}},$the initial estimate maintains the same phase, but the averaged estimatemay not. From this point on, the averaging operation will be referred toas a “smoothing” operation because the values of the resulting“smoothed” estimates are closer in value than the initial estimates.Additionally, the smoothing circuit 120 may be referred to as a“smoothing filter.” The smoothing operation

$\frac{{\hat{h}}_{i} + {\hat{h}}_{i + 1}}{2}$is exemplary and other, more complex smoothing operations may be useddepending on the circumstances. For example, in one embodiment, thesmoothing operation can perform a weighted average. In anotherembodiment, the noise mitigation may be improved by averaging theinitial estimates of a sub-channel and both of its adjacentsub-channels. In one embodiment, the averaging may involve more thanthree sub-channels.

There may be circumstances where the noise between sub-channels are notindependent and are correlated, and the channel effects of adjacentsub-channels are perfectly or highly correlated at least in phase. Inaccordance with one aspect of the invention, the phase adjustmentcircuit 118 may be able to identify and compensate for noise correlationby averaging the phase differences between the initial estimates ofadjacent sub-channels. Specifically, as shown in FIG. 4, unless thenoise effect in a sub-channel occurs in the same direction or theopposite direction as h_(i)·s_(i), the noise effect will adjust thephase of h_(i)·s_(i). If the noise effects in adjacent sub-channels areindependent, the phase adjustments between the adjacent sub-channelswill appear to be random so that the phase difference between theinitial estimates of adjacent sub-channels will also appear to berandom. Accordingly, if the noise effects in adjacent sub-channels areindependent, the average of the phase differences between adjacentsub-channels should come out to be substantially zero. However, if thenoise effects in the sub-channels are slightly correlated, the averageof the phase differences between the initial estimates for adjacentsub-channels should come out to be non-zero.

In one embodiment, the phase adjustment circuit 118 computes the resultof the operation

${\overset{\_}{\Delta\;\varphi} = {\frac{1}{N - 1}{\sum\limits_{i = 0}^{N - 2}\left( {{\arg\left( {\hat{h}}_{i} \right)} - {\arg\left( {\hat{h}}_{i + 1} \right)}} \right)}}},$where N is the number of sub-channels to average, and ĥ_(i), i=0, . . ., (N−1), are the initial estimates of the frequency response of thesub-channels. The value N may be the total number of sub-channels or thenumber of sub-channels in a subset. If the noise effects areuncorrelated, the result Δφ will be substantially zero. Coincidentally,the result Δφ will also be substantially zero if the noise effects areperfectly correlated. When the noise effects in the sub-channels areotherwise correlated, the result Δφ will be non-zero and will representan estimate of the direction of the noise effect in a sub-channel. Inone embodiment, the phase adjustment circuit 118 can compensate thephases of the initial estimates h by computing the result of theoperation ĝ_(i)=ĥ_(i)e^(j(− Δφ)). It can be seen that this operation isharmless if the noise effects are uncorrelated because the value of Δφwill be substantially zero, and there will be no change in the phase ofthe initial estimates. The smoothing circuit 120 can perform itssmoothing operation on the phase-adjusted estimates ĝ_(i), in the samemanner described above, to produce smoothed phase-adjusted estimatesĝ_(i,smoothed). In one embodiment, the smoothed phase-adjusted estimatesĝ_(i,smoothed) are the final estimates of the frequency response of theFDM channel.

In certain circumstances, the channel effects of adjacent sub-channelsmay be correlated but may not be perfectly correlated. In one situationknown as “phase roll,” the channel's phase response may increaselinearly with frequency due to, for example, timing offset at thereceiver. In this situation, the amount of the phase roll can beestimated by the phase adjustment circuit 118 by computing the result ofthe operation

$\overset{\_}{\Delta\;\varphi} = {\frac{1}{N - 1}{\sum\limits_{i = 0}^{N - 2}{\left( {{\arg\left( {\hat{h}}_{i} \right)} - {\arg\left( {\hat{h}}_{i + 1} \right)}} \right).}}}$Assuming the noise effects across different sub-channels areindependent, the result Δφ should approximate the amount of the phaseroll. In this situation, the phase adjustment circuit 118 can compensatethe phases of the initial estimates ĥ_(i) so that the phase adjustmenteffects of the noise can be isolated from the phase roll. The phaseadjustment circuit 118 can compensate the phases of the initialestimates ĥ_(i) by computing the result of the operationĝ_(i)=ĥ_(i)e^(j(−i· Δφ)), for i=0, . . . , (N−1). The phase adjustmentcircuit 118 provides greater compensation at higher values of i becausethe phase roll increases linearly with frequency. The smoothing circuit120 can perform its smoothing operation on the phase-adjusted estimatesĝ_(i) to produce smoothed phase-adjusted estimates ĝ_(i,smoothed). Thesmoothing operation may mitigate the noise effects in the sub-channels,as described above. However, in this situation, the phase-adjustedestimates ĝ_(i,smoothed) are not the final channel estimates because thecorrect channel estimates should include phase roll. Accordingly, inaccordance with one aspect of the invention, the reverse phaseadjustment circuit 122 of the channel estimation circuit canre-introduce the phase roll back into the channel estimates by computingthe result of the operation) ĥ_(i,final)=ĝ_(i,smoothed)e^(j(+i· Δφ)).

The two scenarios illustrated above involve noise correlation in onescenario and phase roll in the other scenario. One difference betweenthe phase adjustment operations in the two scenarios is in whether ornot to perform the reverse phase adjustment operation at the end of theoperation. In certain applications and circumstances, it may not bepossible to predict which scenario is occurring. For such applicationsand circumstances, it may be necessary to simply assume that oneparticular scenario is occurring and to perform the phase adjustmentoperation appropriate for that particular scenario.

Accordingly what have been described thus far are a method and a circuitfor estimating the frequency response of a frequency divisionmultiplexed channel. FIG. 5 shows a flow diagram of one embodiment ofthe disclosed method. In accordance with one aspect of the invention, atransmitter can send a predetermined training signal to a receiver,which can receive a signal that corresponds to a predetermined trainingsignal 502. A channel estimation circuit of the receiver can computeinitial estimates of the frequency response of sub-channels in the FDMchannel 504. The initial estimates may be inaccurate because of noiseeffects in the sub-channels. A phase adjustment circuit can compute amean phase difference 506 and can adjust the phase of the initialestimates based on the mean phase difference 508. The amount of theadjustment is different depending on whether or not there is phase roll.A smoothing circuit can apply a smoothing operation to thephase-adjusted estimates 510. If it is determined that there is phaseroll, a reverse phase adjustment circuit can reintroduce the phase rollback into the smoothed phase-adjusted estimates to provide final channelestimates 512.

In accordance with one aspect of the invention, the disclosed technologycan be deployed in a single input single output (SISO) system, or in amultiple input multiple output (MIMO) system. In a MIMO system, thedisclosed technology can be applied to paired transmitters andreceivers.

In accordance with one aspect of the invention, and with reference toFIG. 1, a receiver 104 can include a control circuit 114 that controlsthe phase adjustment circuit 118, the smoothing circuit 120, and thereverse phase adjustment circuit 122. The transmitter 102 and thereceiver 104 may have multiple modes of operation, and some modes ofoperation may not be amenable to a smoothing operation. For example, inmodes where beam forming or cyclic delay diversity is involved or wherethe transmitter introduces timing offset and phase roll, the controlcircuit 114 can disable the phase adjustment circuit 118, the smoothingcircuit 120, and the reverse phase adjustment circuit 122. In suchmodes, the initial estimation circuit 116 alone can provide the channelestimate.

Referring now to FIGS. 6A-6E, various exemplary implementations of thepresent invention are shown.

Referring now to FIG. 6A, the present invention can be implemented in ahigh definition television (HDTV) 1020. The present invention mayimplement either or both signal processing and/or control circuits,which are generally identified in FIG. 6A at 1022, a WLAN interfaceand/or mass data storage of the HDTV 1020. The HDTV 1020 receives HDTVinput signals in either a wired or wireless format and generates HDTVoutput signals for a display 1026. In some implementations, signalprocessing circuit and/or control circuit 1022 and/or other circuits(not shown) of the HDTV 1020 may process data, perform coding and/orencryption, perform calculations, format data and/or perform any othertype of HDTV processing that may be required.

The HDTV 1020 may communicate with mass data storage 1027 that storesdata in a nonvolatile manner such as optical and/or magnetic storagedevices. The HDD may be a mini HDD that includes one or more plattershaving a diameter that is smaller than approximately 1.8″. The HDTV 1020may be connected to memory 1028 such as RAM, ROM, low latencynonvolatile memory such as flash memory and/or other suitable electronicdata storage. The HDTV 1020 also may support connections with a WLAN viaa WLAN network interface 1029.

Referring now to FIG. 6B, the present invention implements a controlsystem of a vehicle 1030, a WLAN interface and/or mass data storage ofthe vehicle control system. In some implementations, the presentinvention may implement a powertrain control system 1032 that receivesinputs from one or more sensors such as temperature sensors, pressuresensors, rotational sensors, airflow sensors and/or any other suitablesensors and/or that generates one or more output control signals such asengine operating parameters, transmission operating parameters, and/orother control signals.

The present invention may also be implemented in other control systems1040 of the vehicle 1030. The control system 1040 may likewise receivesignals from input sensors 1042 and/or output control signals to one ormore output devices 1044. In some implementations, the control system1040 may be part of an anti-lock braking system (ABS), a navigationsystem, a telematics system, a vehicle telematics system, a lanedeparture system, an adaptive cruise control system, a vehicleentertainment system such as a stereo, DVD, compact disc and the like.Still other implementations are contemplated.

The powertrain control system 1032 may communicate with mass datastorage 1046 that stores data in a nonvolatile manner. The mass datastorage 1046 may include optical and/or magnetic storage devices forexample hard disk drives HDD and/or DVDs. The HDD may be a mini HDD thatincludes one or more platters having a diameter that is smaller thanapproximately 1.8″. The powertrain control system 1032 may be connectedto memory 1047 such as RAM, ROM, low latency nonvolatile memory such asflash memory and/or other suitable electronic data storage. Thepowertrain control system 1032 also may support connections with a WLANvia a WLAN network interface 1048. The control system 1040 may alsoinclude mass data storage, memory and/or a WLAN interface (all notshown).

Referring now to FIG. 6C, the present invention can be implemented in acellular phone 1050 that may include a cellular antenna 1051. Thepresent invention may implement either or both signal processing and/orcontrol circuits, which are generally identified in FIG. 6C at 1052, aWLAN interface and/or mass data storage of the cellular phone 1050. Insome implementations, the cellular phone 1050 includes a microphone1056, an audio output 1058 such as a speaker and/or audio output jack, adisplay 1060 and/or an input device 1062 such as a keypad, pointingdevice, voice actuation and/or other input device. The signal processingand/or control circuits 1052 and/or other circuits (not shown) in thecellular phone 1050 may process data, perform coding and/or encryption,perform calculations, format data and/or perform other cellular phonefunctions.

The cellular phone 1050 may communicate with mass data storage 1064 thatstores data in a nonvolatile manner such as optical and/or magneticstorage devices for example hard disk drives HDD and/or DVDs. The HDDmay be a mini HDD that includes one or more platters having a diameterthat is smaller than approximately 1.8″. The cellular phone 1050 may beconnected to memory 1066 such as RAM, ROM, low latency nonvolatilememory such as flash memory and/or other suitable electronic datastorage. The cellular phone 1050 also may support connections with aWLAN via a WLAN network interface 1068.

Referring now to FIG. 6D, the present invention can be implemented in aset top box 1080. The present invention may implement either or bothsignal processing and/or control circuits, which are generallyidentified in FIG. 6D at 1084, a WLAN interface and/or mass data storageof the set top box 1080. The set top box 1080 receives signals from asource such as a broadband source and outputs standard and/or highdefinition audio/video signals suitable for a display 1088 such as atelevision and/or monitor and/or other video and/or audio outputdevices. The signal processing and/or control circuits 1084 and/or othercircuits (not shown) of the set top box 1080 may process data, performcoding and/or encryption, perform calculations, format data and/orperform any other set top box function.

The set top box 1080 may communicate with mass data storage 1090 thatstores data in a nonvolatile manner. The mass data storage 1090 mayinclude optical and/or magnetic storage devices for example hard diskdrives HDD and/or DVDs. The HDD may be a mini HDD that includes one ormore platters having a diameter that is smaller than approximately 1.8″.The set top box 1080 may be connected to memory 1094 such as RAM, ROM,low latency nonvolatile memory such as flash memory and/or othersuitable electronic data storage. The set top box 1080 also may supportconnections with a WLAN via a WLAN network interface 1096.

Referring now to FIG. 6E, the present invention can be implemented in amedia player 1100. The present invention may implement either or bothsignal processing and/or control circuits, which are generallyidentified in FIG. 6E at 1104, a WLAN interface and/or mass data storageof the media player 1100. In some implementations, the media player 1100includes a display 1107 and/or a user input 1108 such as a keypad,touchpad and the like. In some implementations, the media player 1100may employ a graphical user interface (GUI) that typically employsmenus, drop down menus, icons and/or a point-and-click interface via thedisplay 1107 and/or user input 1108. The media player 1100 furtherincludes an audio output 1109 such as a speaker and/or audio outputjack. The signal processing and/or control circuits 1104 and/or othercircuits (not shown) of the media player 1100 may process data, performcoding and/or encryption, perform calculations, format data and/orperform any other media player function.

The media player 1100 may communicate with mass data storage 1110 thatstores data such as compressed audio and/or video content in anonvolatile manner. In some implementations, the compressed audio filesinclude files that are compliant with MP3 format or other suitablecompressed audio and/or video formats. The mass data storage may includeoptical and/or magnetic storage devices for example hard disk drives HDDand/or DVDs. The HDD may be a mini HDD that includes one or moreplatters having a diameter that is smaller than approximately 1.8″. Themedia player 1100 may be connected to memory 1114 such as RAM, ROM, lowlatency nonvolatile memory such as flash memory and/or other suitableelectronic data storage. The media player 1100 also may supportconnections with a WLAN via a WLAN network interface 1116. Still otherimplementations in addition to those described above are contemplated.

Accordingly, what have been described herein are a method and apparatusfor estimating the frequency response of a FDM channel. The disclosedmethods, components, and circuits can be implemented using variousanalog and/or digital circuit means, including circuitry made fromvarious types, sizes, and/or configurations of transistors, MOStransistors, field effect transistors, BJTs, diodes, resistors,capacitors, inductors, integrated circuits, operation amplifiers,operational transconductance amplifiers, comparators, registers,latches, and/or current sources. The disclosed methods and systems canalso be implemented using a processor architecture having machinereadable instructions. The disclosed embodiments and illustrations areexemplary and do not limit the scope of the disclosed invention asdefined by the following claims.

1. A method for estimating a frequency response of a channel, the methodcomprising: adjusting phase components of estimates of the frequencyresponse to provide phase-adjusted estimates; performing a smoothingoperation on the phase-adjusted estimates to provide smoothedphase-adjusted estimates of the frequency response of the channel; andreversing the phase adjustment of at least one of the adjusted phasecomponents, wherein the reversing of the phase adjustment is performedon the smoothed phase-adjusted estimates of the frequency response ofthe channel.
 2. The method of claim 1, wherein adjusting the phasecomponents of the estimates comprises: for each of at least one ofsub-channels of the channel, computing a phase difference between aphase component of an estimate for the sub-channel and a phase componentof an estimate for an adjacent sub-channel; computing a mean phasedifference based on the phase differences; and adjusting the phasecomponents of the estimates based on the mean phase difference toprovide the phase-adjusted estimates.
 3. The method of claim 2, whereinthe sub-channels include at least a first sub-channel and a secondsub-channel, and wherein adjusting the phase components of the estimatesbased on the mean phase difference comprises: adjusting the phasecomponent of the estimate for the first sub-channel by a first multipleof the mean phase difference; and adjusting the phase component of theestimate for the second sub-channel by a second multiple of the meanphase difference.
 4. The method of claim 3, wherein the firstsub-channel corresponds to a first frequency value and the secondsub-channel corresponds to a second frequency value, and wherein: thefirst multiple is greater than the second multiple if the firstfrequency value is greater than the second frequency value; and thefirst multiple is less than the second multiple if the first frequencyvalue is less than the second frequency value.
 5. The method of claim 3,wherein the first multiple differs from the second multiple by an amountthat is proportional to a difference between the first and secondfrequency values.
 6. The method of claim 2, wherein the reversingcomprises: for each sub-channel, adjusting a phase component of thesmoothed phase-adjusted estimate by an amount that is substantiallyopposite of an amount of phase adjusting of the estimate for thesub-channel.
 7. The method of claim 1, wherein performing the smoothingoperation on the phase-adjusted estimates comprises: for each ofsub-channels of the channel, computing the smoothed phase-adjustedestimate for the sub-channel based on the phase-adjusted estimate forthe sub-channel and at least the phase-adjusted estimate for an adjacentsub-channel.
 8. The method of claim 7, wherein computing the smoothedphase-adjusted estimate for the sub-channel comprises computing anaverage of the phase-adjusted estimate for the sub-channel and thephase-adjusted estimates for a particular number of neighboringsub-channels.
 9. The method of claim 1, further comprising: accessing anindicator that has an on-value and at least one other value; andperforming said adjusting, said smoothing, and said reversing only whenthe indicator has the on-value.
 10. The method of claim 1, furthercomprising computing the estimates of the frequency response ofsub-channels of the channel based on a received signal.
 11. An apparatusfor estimating a frequency response of a channel, the apparatuscomprising: a phase adjustment circuit configured to adjust phasecomponents of estimates of the frequency response to providephase-adjusted estimates; a smoothing circuit configured to perform asmoothing operation on the phase-adjusted estimates to provide smoothedphase-adjusted estimates of the frequency response of the channel; and areverse phase adjustment circuit configured to reverse the adjustment ofat least one of the adjusted phase components in the smoothedphase-adjusted estimates of the frequency response of the channel. 12.The apparatus of claim 11, wherein the phase adjustment circuitcomprises: a phase difference circuit configured to compute, for each ofat least one of sub-channels of the channel, a phase difference betweeni) a phase component of an estimate for the sub-channel, and ii) a phasecomponent of an estimate for an adjacent sub-channel; an averagingcircuit configured to compute a mean phase difference based on the phasedifferences; and an adjustment circuit configured to compute the phasecomponents of the estimates based on the mean phase difference toprovide the phase-adjusted estimates.
 13. The apparatus of claim 12,wherein the sub-channels include at least a first sub-channel and asecond sub-channel, and wherein the adjustment circuit comprises:circuitry configured to adjust the phase component of the estimate forthe first sub-channel by a first multiple of the mean phase difference;and circuitry configured to adjust the phase component of the estimatefor the second sub-channel by a second multiple of the mean phasedifference.
 14. The apparatus of claim 13, wherein the first sub-channelcorresponds to a first frequency value and the second sub-channelcorresponds to a second frequency value, and wherein: the first multipleis greater than the second multiple if the first frequency value isgreater than the second frequency value, and the first multiple is lessthan the second multiple if the first frequency value is less than thesecond frequency value.
 15. The apparatus of claim 13, wherein the firstmultiple differs from the second multiple by an amount that isproportional to a difference between the first and second frequencyvalues.
 16. The apparatus of claim 12, wherein the reverse phaseadjustment circuit is configured to adjust, for each sub-channel, aphase component of the smoothed phase-adjusted estimate by an amountthat is substantially opposite of an amount of phase adjusting of theestimate for the sub-channel.
 17. The apparatus of claim 11, wherein thesmoothing circuit is configured to compute, for each of sub-channels ofthe channel, the smoothed phase-adjusted estimate for the sub-channelbased on i) the phase-adjusted estimate for the sub-channel and ii) atleast the phase-adjusted estimate for an adjacent sub-channel.
 18. Theapparatus of claim 11, further comprising: a memory configured to storean indicator that has an on-value and at least one other value; and acontrol circuit configured to enable the phase adjustment circuit, thesmoothing circuit, and the reverse phase adjustment circuit only whenthe indicator has the on-value.
 19. The apparatus of claim 11, furthercomprising an estimation circuit configured to compute estimates of thefrequency response of sub-channels of the channel based on a receivedsignal.
 20. A method for estimating a frequency response of a channel,the method comprising: compensating phase components of estimates of thefrequency response by removing a corresponding phase roll from at leastone of the phase components to provide phase-adjusted estimates of thefrequency response of the channel; performing a smoothing operation onthe phase-adjusted estimates to provide smoothed phase-adjustedestimates of the frequency response; and re-introducing the respectivephase roll to the at least one of the phase components in the smoothedphase-adjusted estimates of the frequency response.